Ac-dc power converter with power factor correction

ABSTRACT

The present invention relates to an AC-DC power converter which comprises a resonant DC-DC converter and a charge pump circuit. The charge pump circuit is configured to perform power factor correction of the AC-DC power converter by drawing current pulses at a switching frequency of the converter from an AC line voltage such that electrical charges of the current pulses vary substantially proportionally with instantaneous amplitude of the AC line voltage.

FIELD OF THE INVENTION

The present invention relates to an AC-DC power converter whichcomprises a resonant DC-DC converter and a charge pump circuit. Thecharge pump circuit is configured to perform power factor correction ofthe converter by drawing current pulses at a switching frequency of theconverter from an AC line voltage such that electrical charges of thecurrent pulses vary substantially proportionally to instantaneousamplitude of the AC line voltage.

BACKGROUND OF THE INVENTION

The present AC-DC power converters may be applied to power factorcorrecting AC-DC power converters in numerous applications. A high powerfactor is generally required or at least highly desirable in a powerdelivery system to reduce power losses and reduce distortion introducedto the AC grid by input current harmonics. In case the power system isloaded by a nonlinear load, e.g. switching converter, current drawn bythe load is interrupted by a switching activity and therefore containsnumerous higher frequency components that are multiples of the powersystem frequency. Such harmonic distortion reduces average powertransferred to a load of the AC-DC power converter in addition tocontaminating the AC grid. Power factor correction brings the powerfactor of a power circuit closer to 1 by making the load appear moreresistive to the AC grid. Thus, achieving a close-to-sinusoidal linecurrent that is substantially proportional to, and substantiallyin-phase with, the AC grid-voltage.

The skilled person will understand that the AC-DC power convertersdisclosed herein may be utilized in a wide range of applications andproduct categories, including laptop chargers, LED drivers, otheradapters and power supplies for various industrial and consumerelectronics. Thus, the AC-DC power converters disclosed herein meet anincreasing demand for smaller, more power or energy efficient and longerliving AC-DC converters. The present AC-DC power converter design solvesthese and other problems by a charge pump based Power Factor Correction(PFC). The present AC-DC converter may for example use soft-switchinginverter topologies and state-of-the-art devices such as wide bandgapsemiconductors and relevant magnetic materials as discussed below inadditional detail.

With the current trend towards smaller and highly portable electronicequipment for consumer and industrial applications it is important tominimize weight and size of the equipment without sacrificingperformance. One application of the present AC-DC converter with a greatdemand for miniaturization is offline power converters. Conventionaloffline converters are 2-stage architectures wherein the first stage isan AC-DC power factor correction (PFC) rectifier followed by an energystorage capacitor to filter the double-the-line frequency component onthe output, while the second stage is a DC-DC converter providing thevoltage and current levels conforms to load electrical characteristics.This conversion has to comply with a number of regulations dictating theshape of the input current to limit the mains voltage distortion [1],[2].

Pulse-width-modulated (PWM) converters have been the primary candidatefor the AC-DC stage in offline converters, including step-downconverters, step-up converters, buck-boost converters, flybackconverters and single-ended primary-inductor converter (SEPIC). Theseprior art converter topologies can provide high power factor, butgenerally suffer from several pronounced problems such as high conductedelectromagnetic interference (EMI) from rectangular switching waveforms.Another problem associated with prior art converter topologies ishard-switching operation which typically leads to severe switchinglosses, low overall energy efficiency, severe EMI problems, relativelylow switching frequencies and voltage stress, dv/dt, problems etc.

On the other hand, resonant inverters and converters typically havesubstantially lower switching losses than their PWM counterparts. Theresonant inverters of AC-DC power converters in accordance with thepresent invention may therefore be configured to providezero-voltage-switching (ZVS) and/or zero-current-switching (ZCS) toprovide high power conversion efficiencies at high switching frequenciesfor example switching frequencies above 750 kHz or above 1 MHz. That inturn results in reduced sizes of passive electrical components and thushigher power densities, higher loop-gain bandwidths and faster transientresponses. The resonant inverters and converters typically allowincorporation of high frequency transformers in the resonant tank orcircuit. The latter allows galvanic isolation, in addition to offeringdifferent output voltages. That, in turn, can help combine the AC-DCwith the following DC-DC stage for a single-stage solution for thepresent offline converters.

SUMMARY OF THE INVENTION

A first aspect of the invention relates to an AC-DC power converterwhich comprises an AC rectification circuit configured to convert an ACline voltage into a rectified line voltage and a rectifying elementconnected between the rectified line voltage and a DC supply voltage ofa resonant DC converter. The resonant DC converter comprises a resonantinverter configured to convert the DC supply voltage into a resonantinverter voltage at a fixed or controllable switching frequency and anoutput rectification circuit configured to generate a DC output voltagefrom the resonant voltage or generate a DC output current from theresonant voltage for supply to a converter load such as an LED lampassembly. The AC-DC power converter additionally comprises a charge pumpcircuit connected to the rectified line voltage and the resonantinverter voltage. The charge pump circuit is configured to draw currentpulses at the fixed or controllable switching frequency from the AC linevoltage wherein electrical charges of the current pulses varyproportionally to, or at least substantially proportionality to, aninstantaneous amplitude of the AC line voltage.

The skilled person will understand that the charge pump circuit mayoperate in an open loop manner and therefore does not require anyseparate regulation loop or mechanism to carry out the power factorcorrection of the line current drawn from the AC line voltage or mainsvoltage. Hence, the AC-DC power converter may comprise merely a singlefeedback regulation loop to adjust the DC output voltage or DC outputcurrent despite the integrated power factor correction (PFC) mechanismprovided by the charge pump circuit. The AC-DC power converter may forexample comprise a voltage or current regulation loop configured toadjust the DC output voltage or DC output current in accordance with aDC reference voltage or a DC reference current. The adjustment of the DCoutput voltage or DC output current may be achieved by controlling theswitching frequency of the resonant inverter, which is also theswitching frequency of the resonant DC-DC converter, as discussed inadditional detail below with reference to the appended drawings.

Other embodiments of the AC-DC power converter may have a substantiallyfixed switching frequency and the adjustment of the DC output voltage orDC output current may be achieved by duty cycle control, i.e. on/offcontrol, of the switch control signal which may be PWM modulated.

The resonant inverter may comprise a series resonant network or tank, aparallel resonant network or tank or a combination of both. In eachcase, the resonant network or tank may comprise at least an inductor anda capacitor to set a resonance frequency. The resonance frequency maylie between 100 kHz and 300 MHz, for example above 750 kHz, and tuned tocoincide with the switching frequency of the resonant DC-DC converter,which therefore also may lie between 100 kHz and 300 MHz. The skilledperson will understand that properties of active components of the AC-DCpower converter, such as transistors, and passive components, such asthe inductor and capacitor of the resonant network, to may be tailoredto a specific switching frequency or switching frequency range.

The skilled person will understand that the resonant inverter may haveany of numerous well-known topologies such as a topology selected fromthe group {class E, class F, class DE, class EF, LLC}. Some embodimentsof the AC-DC power converter may comprise a galvanic isolation barrier,e.g. a transformer with a certain conversion ratio, to step-up orstep-down the resonant inverter voltage. The transformer may providegalvanic isolation between primary side circuitry and secondary sidecircuitry of the AC-DC power converter. The galvanic isolation barriermay be coupled between the resonant output voltage and an input voltageof the output rectification circuit.

According to one embodiment of the AC-DC power converter, the chargepump circuit comprises a smoothing capacitor connected to the DC supplyvoltage of the resonant inverter and a pump capacitor, or flyingcapacitor, connected from the resonant inverter voltage to the rectifiedline voltage. The circuit topology of the AC-DC power converter ensuresthat the charge variation of the pump capacitor, which is proportionalto the voltage variation across the pump capacitor, follows the AC linevoltage across the 50/60 Hz line cycle. Accordingly, the average inputor line current drawn by the AC-DC power converter substantially followsthe AC line voltage and a unity power factor can ideally be obtainedeven though minor component and circuit imperfections may leave thepower factor slightly below unity. The charge pump circuit may beconfigured such that the proportionality between the electrical chargesof the current pulses and the instantaneous amplitude of the AC linevoltage leads to a power factor exceeding 0.95, and more preferablyexceeds 0.98, for example exceeding 0.99 as evidenced by theexperimental results discussed in detail below with reference to theappended drawings.

According to certain embodiments of the invention, the electricalinterconnection between the rectified line voltage and resonant invertervoltage is based on a capacitor only network, for example exclusively bythe pump capacitor. This provides a compact and low-cost connection thatmay be based on standard, low cost and readily available capacitors. Theelectrical interconnection may therefore avoid the use of an inductivecoupling or an inductive component of the below-discussed galvanicisolation transformer, such as a separate transformer winding, back tothe rectified line voltage.

The charge pump circuit may be configured to, during a cycle of theswitching frequency, sequentially cycle through states of:

-   -   a first state where each of the rectifying element (D_(p)) and        AC rectification circuit (D_(B)) is non-conducting/off and a        voltage across the pump/flying capacitor (C_(P)) remains        substantially constant;    -   a second state where AC rectification circuit (D_(B)) is        conducting/on and the rectifying element (D_(p)) is        non-conducting/off to charge the pump/flying capacitor (C_(P))        by line current drawn from the AC line voltage;    -   a third state where each of the rectifying element (D_(p)) and        AC rectification circuit (D_(B)) is non-conducting/off and the        voltage across the pump/flying capacitor (C_(P)) remains        substantially constant;    -   a fourth state where the AC rectification circuit (D_(B)) is in        a non-conducting/off state and the rectifying element (D_(p)) is        in a conducting/on state such that discharge current flows from        the pump/flying capacitor (C_(P)) into the smoothing capacitor        (C_(DC)) to increase the DC supply voltage (V_(DC)) of the        resonant inverter and decrease the voltage across the        pump/flying capacitor (C_(P)).

The charge pump circuit may be configured to cycle through its secondstate during a rising edge of a waveform of the resonant invertervoltage; and cycle through its fourth state during a falling edge of thewaveform of the resonant inverter voltage as discussed in additionaldetail below with reference to the appended drawings.

According to one embodiment, a capacitance of the smoothing capacitorand a capacitance of the pump/flying capacitor are selected such thatthe DC supply voltage of the resonant inverter is higher than the ACline voltage across every cycle of the AC line voltage. An advantage ofthe latter selection of respective capacitances of the smoothingcapacitor and pump capacitor is that it prevents cross conductionbetween AC line voltage source and the DC supply voltage (V_(DC)) of theresonant DC-DC converter.

The AC-DC power converter preferably comprises a voltage regulation loopor current regulation loop configured to adjust the DC output voltage(V_(OUT)) or DC output current, respectively, in accordance with a DCreference voltage or a DC reference current. The voltage or currentregulation loop may be configured to:

-   -   adjust the DC output voltage or DC output current by adjusting        or controlling the switching frequency, i.e. frequency        modulation control, and/or    -   adjust the DC output voltage (V_(OUT)) or DC output current by        off/on modulation or duty cycle modulation of the switching        frequency. Hence, in the latter embodiment, the switching        frequency of AC-DC power converter may be fixed and the AC-DC        power converter turned-on and turned-off at a certain control        frequency.

According to one embodiment, the voltage or current regulation loop maybe configured to adjust the switching frequency of the AC-DC powerconverter with more than +/−5%, or even more than +/−10%, relative to anominal switching frequency of the AC-DC power converter. The nominalswitching frequency may be identical to the resonance frequency of theresonant network or a predetermined off-set relative to resonancefrequency of the resonant network.

The resonant inverter preferably comprises at least one semiconductorswitch connected between the DC supply voltage (V_(DC)) and a negativesupply rail. The at least one semiconductor switch may comprise one ormore wide bandgap transistors such as one or more gallium nitrideFET(s). The resonant inverter may comprise controllable switch networksuch as a half-bridge driver comprising a pair of wide bandgaptransistors. A switch signal, at the fixed or controllable switchingfrequency, may be applied to a control terminal, e.g. a gate terminal,of the least one semiconductor switch of the resonant inverter. Anoutput terminal of the controllable switch network may be connected to afirst end of the resonant network.

The resonant inverter may comprise a controllable switch networkexhibiting a topology selected from a group of: class DE, Class E, classEF, LLC.

The resonant inverter may be configured for zero voltage switching (ZVS)and/or zero current switching (ZCS). The output rectification circuit ofthe DC-DC converter may be configured for zero voltage switching (ZVS)and/or zero current switching (ZCS).

The output rectification circuit of the DC-DC converter may comprise oneor more passive diodes, such as silicon carbide Schottky diode(s), orone more active/controllable diodes such as one or more transistors suchas at least one MOSFET.

A second aspect of the invention relates to a method of applying powerfactor correction to an AC-DC power converter using a charge pumpcircuit, said method comprising steps of:

-   -   converting an AC line voltage into a rectified line voltage        (V_(B));    -   applying the rectified line voltage (V_(B)) to a DC supply        voltage (V_(DC)) of a resonant DC-DC converter through a        rectifying element (D_(p)), such as a semiconductor diode;    -   generating a resonant inverter voltage by switching a resonant        inverter at a fixed or controllable switching frequency;    -   rectifying the resonant inverter voltage to generate a DC output        voltage or generate a DC output current;    -   drawing charging pulses, at the switching frequency, from the AC        line voltage into a pump or flying capacitor (C_(P)) connected        between the rectified line voltage (V_(B)) and the resonant        inverter voltage, wherein electrical charges of the charging        pulses vary substantially proportionally to an instantaneous        amplitude of the AC line voltage;    -   discharging the pump or flying capacitor into a smoothing        capacitor, connected to the DC supply voltage, by supplying        current pulses, at the switching frequency, into the smoothing        capacitor.

The skilled person will understand that present methodology may comprisethat the charge pump circuit sequentially cycles through the previouslydiscussed first, second, third and fourth states during every cycle ofthe switching frequency.

BRIEF DESCRIPTION OF THE DRAWINGS

A preferred embodiment of the invention will be described in more detailin connection with the appended drawings, in which:

FIG. 1 is a top-level block diagram of AC-DC power converters inaccordance with the invention,

FIG. 2 is a simplified electrical circuit diagram of a class-DE seriesresonant converter of the AC-DC power converter,

FIG. 2A shows block diagrams of various exemplary embodiments of AC-DCpower converters in accordance with the invention,

FIG. 3 is a simplified equivalent diagram of the charge pump circuit ofthe AC-DC power converter,

FIG. 4 is a simplified electrical diagram of a first exemplaryembodiment of the AC-DC power converter based on a class-DE converterand a charge-pump circuit,

FIG. 4A is a simplified electrical diagram of a second exemplaryembodiment of the AC-DC power converter based on a class-DE converterand a charge-pump circuit,

FIG. 5 shows low-frequency operation of the charge pump circuit across ahalf-line cycle, 50 Hz, of the mains line voltage inputted to the AC-DCpower converter,

FIG. 6 illustrates high frequency operation of an exemplary embodimentof the AC-DC power converter across two switching cycles of theswitching frequency of the class DE resonant converter,

FIG. 7 shows simulation results for the mains line current and mainsvoltage input of the exemplary embodiment of the AC-DC power converter,

FIG. 8 shows simulated waveforms of various internal voltages andcurrents of the exemplary embodiment of the AC-DC power converter,

FIG. 9 shows simulated losses of an exemplary inductor of a resonantnetwork of the class-DE converter,

FIG. 10 simulated core losses at 1 MHz of the exemplary inductor of theresonant network of the class-DE converter

FIG. 11—upper plot shows measured output power and conversion efficiencyof an experimental prototype AC-DC power converter across itsoperational frequency range from 960 kHz to 1040 kHz,

FIG. 11—lower plot shows measured PFC results of the experimentalprototype AC-DC power converter across its operational frequency rangefrom 960 kHz to 1040 kHz,

FIG. 12 shows measured harmonics distribution of the mains line currentat full-load operation and half-load operation of the experimentalprototype AC-DC power converter,

FIG. 13 shows line-frequency time domain waveforms of the experimentalprototype AC-DC power converter; and

FIG. 14 shows measured harmonics distribution of the mains line currentfor the experimental prototype AC-DC power converter at 120 V_RMS and230 V_RMS line voltage inputs.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

In the following, various exemplary embodiments of the present AC-DCpower converter are described with reference to the appended drawings.The skilled person will understand that the accompanying drawings areschematic and simplified for clarity and therefore merely show detailswhich are essential to the understanding of the invention, while otherdetails have been left out. Like reference numerals refer to likeelements or components throughout. Like elements or components willtherefore not necessarily be described in detail with respect to eachfigure. It will further be appreciated that certain actions and/or stepsmay be described or depicted in a particular order of occurrence whilethose skilled in the art will understand that such specificity withrespect to sequence is not actually required.

FIG. 1 shows a simplified block diagram of the present AC/DC powerconverter 100. The AC/DC power converter 100 comprises an ACrectification circuit (101) which is coupled to an AC line voltage(V_(IN)) which may deliver AC mains voltages like 230 V/50 Hz or 110V/60Hz. The AC rectification circuit (101) may include a mains input filteras illustrated below. The AC rectification circuit (101) is configuredto convert the AC line voltage V_(IN) into a rectified line voltage(V_(B)). The AC/DC power converter 100 additionally comprises a chargepump circuit (103), which preferably included a smoothing capacitor(C_(DC)), and class DE resonant converter (105). The smoothing capacitor(C_(DC)) is connected to the DC supply voltage (V_(DC)) of the resonantinverter (101). The resonant converter (105) is configured to convertthe DC supply voltage (V_(DC)) firstly into a resonant inverter voltage(illustrated below) at a fixed or adjustable switching frequency.Secondly, an output rectification circuit (not shown) of the resonantconverter (105) generates a DC output voltage (V_(OUT)) of the AC/DCpower converter 100 by rectifying the resonant inverter voltage. Aschematically illustrated converter load (R_(L)), such as a LED lampassembly, is connected to the DC output voltage (V_(OUT)). The converterload may generally exhibit inductive, capacitive or resistive impedance.The charge pump circuit (103) is configured to perform power factorcorrection (PFC) of the AC/DC power converter 100 by drawing chargingpulses from the AC line voltage where electrical charges of the chargingpulses vary substantially proportionally with an instantaneous amplitudeof the AC line voltage as discussed in additional detail below.

FIG. 2 shows a simplified electrical circuit diagram of the resonantclass DE power converter (105). The class DE power converter comprises acontrollable switch network 212, a series resonant tank or network (214)comprising an inductor L_(RES) and a capacitor C_(RES). The skilledperson will understand that other types of resonant tanks or tankcircuits (214) may be used in alternative embodiments of the presentAC/DC power converters as schematically illustrated on FIG. 2A. Theclass DE power converter comprises the previously discussed rectifier orrectification circuit 216. The skilled person will appreciate thatalternative resonant power converter topologies such as Class E or LLCtopologies may be employed in the present AC/DC power converter 100. Thecontrollable switch network 212 is connected to positive and negativeterminals or nodes of the DC supply voltage (V_(DC)) to energize theconverter 105. The resonant class DE power converter 105 typicallycomprises of two stages, wherein the first stage comprises thecontrollable switch network 212 and the resonant tank 214. Thecontrollable switch network 212 converts a DC voltage input from the DCsupply voltage (V_(DC)) into a high frequency AC output, i.e. theresonant inverter voltage V_(REC) and the series resonant tank 214 mayperform an AC-AC gain. The second stage of the resonant converter 105comprises the high frequency AC-DC rectification circuit 216 whereinenergy/power supplied by the series resonant tank may be tapped off anddelivered to the converter/output load (R_(L)). The rectificationcircuit 216 may comprise SiC Schottky diodes D_(R1) and D_(R2) asdiscussed in additional detail below. The skilled person will understandthat other types of AC-DC rectification circuits 216 may be used inalternative embodiments of the present AC/DC power converters, includingrectification circuits that include an isolation transformer such as ancenter-tapped isolation transformer as schematically illustrated on FIG.2A.

The utilization of a resonant power converter (105) allows utilizationof soft-switching techniques through the intrinsic alternating behaviourof the currents and voltages through high-side and low-sidesemiconductor switches Q_(HS) and Q_(LS) of the half-bridge 212, orcontrollable switch network, resulting in substantially lower switchinglosses. Driving signals (not shown) to the gate driver 213 arepreferably synchronized with the same duty cycle and extended dead-timeadjustment to avoid cross conduction between the semiconductor switchesQ_(HS) and Q_(LS) switches and allow the resonant inductor current tocharge or discharge the output capacitance of the half-bridge switchesQ_(HS) and Q_(LS) so that their drain voltages reach the appropriatesupply rail voltage before switching the gate. Therefore, ensuring zerovoltage switching (ZVS) operating conditions of the half-bridge 212.

As discussed below in connection with the design of the charge pumpstage, a gain of the class DE resonant inverter or stage shouldpreferably be relatively high, e.g. a gain from about 0.5 to 1 toprovide high power factor and low total harmonic distortion (THD) of theAC input current waveform. The inventors have found that a goodapproximation would be to design for 300 V of DC output voltage andassuming an input DC voltage to the half-bridge equal to the peak of themains input voltage, e.g. 325 V. The design is preferably based on thewell-known First Harmonic Approximation (FHA) approach even thoughalternative procedures may be used. The computation procedure or flowfor an exemplary design of the AC/DC power converter with a targetoutput power of 50 W may start by calculating a rectifier inputresistance R_(REC) from the load resistance R_(L) through impedancetransformation via the resonant rectifier as follows:

$\begin{matrix}{R_{L} = {\frac{{V_{OUT}}^{2}}{P_{OUT}} = {1.8\mspace{11mu} k\;\Omega}}} & (1) \\{R_{REC} = {\frac{2R_{L}}{\pi^{2}} = {365\mspace{11mu}\Omega}}} & (2)\end{matrix}$

The converter voltage conversion ratio is equal to:

M _(V) =V _(OUT) /V _(IN)=0.923  (3)

While gains of the half-bridge 212 and class-DE rectifier are equal to0.45 and 2.22 respectively, the required gain of the series resonanttank 214 can be calculated by:

$\begin{matrix}{M_{V\_ RES} = {\frac{M_{V}}{M_{V\_ HB} \cdot M_{V\_ REC}} = {{0.9}24}}} & (4)\end{matrix}$

The converter loaded quality factor is calculated using the followingequation:

$\begin{matrix}{Q_{L} = \sqrt{\frac{1}{\frac{{M_{V\_ RES}}^{2}}{f_{n} - \frac{1}{f_{n}}}} - 1}} & (5)\end{matrix}$

where f_(n) is the normalized switching frequency, equal tof_(sw)/f_(o)—with f_(o) being a resonant frequency of the seriesresonant tank 214. In order to ensure the validity of the aboveequations given the FHA approach, the loaded quality factor Q_(L) of theresonant tank or circuit 214 needs to be high enough so that theresonant inverter current through the resonant tank 214 is substantiallysinusoidal. A loaded quality factor of 2.5 is chosen for this design[5]. Using equation (5) to calculate the normalized switching frequency,and for a switching frequency of 1 MHz, the resonant frequency iscalculated to 921 kHz. Therefrom, the resonant circuit component valuesmay be calculated as:

$\begin{matrix}{L_{RES} = {\frac{Q_{L} \cdot R_{REC}}{\omega_{o}} = {157\mspace{11mu}\mu\; H}}} & (6) \\{C_{RES} = {\frac{1}{\omega_{o} \cdot Q_{L} \cdot R_{REC}} = {190\mspace{11mu} p\; F}}} & (7)\end{matrix}$

Rectifier devices stresses are calculated as follows:

V _(D_max) =V _(OUT)=300 V  (8)

I _(D_max) =πI _(OUT)=524 mA  (9)

A voltage stress across the half-bridge switches:

V _(S_max) =V _(IN)=325 V  (10)

The current stresses of the half-bridge semiconductor switches Q_(HS)and Q_(LS) and components of the series resonant network may becalculated from:

$\begin{matrix}{I_{RES\_ max} = {\pi{P_{OUT}\left( {\frac{2}{\eta \cdot V_{IN\_ pk}} + \frac{1}{V_{OUT}}} \right)}}} & (11)\end{matrix}$

FIG. 3 shows a simplified electrical equivalent circuit or diagram of anexemplary charge pump circuit 103 of the present AC-DC power converter100. Through the addition of a capacitor, a diode and the previouslydiscussed smoothing capacitor (C_(DC)) to the resonant converter (105),mains input current drawn from the AC line voltage can be regulated tobe substantially proportional to the instantaneous AC line voltage. Thepresent embodiment of the resonant converter (105) comprises aseries-resonant inverter circuit in conjunction with and ahigh-frequency rectification circuit to support LED driver applications.The smoothing capacitor (C_(DC)), or DC energy storage capacitor, isarranged at the converter input. The illustrated electrical equivalentcircuit of the charge pump circuit 103 comprises a pump or flyingcapacitor C_(P) and pump diode D_(P). The AC rectification circuit (101)is schematically represented by diode D_(B) and the rectifier inputvoltage (V_(REC)) is modelled by an independent high frequencysquare-wave voltage source 321. The capacitor C_(DC) is preferablydesigned in accordance with the pump capacitor C_(P), such that the DCsupply voltage V_(DC) for power supply of the resonant inverter isalways higher than the AC line voltage V_(IN) such that the ACrectification circuit D_(B) and the pump diode D_(P) do notcross-conduct. Consequently, the input current I_(IN) drawn from the ACline voltage is equal to a positive charging current I_(P) of the pumpcapacitor C_(P). In other words the capacitance of the smoothingcapacitor (C_(DC)) and the capacitance of the pump/flying capacitor(C_(P)) are selected such that the DC supply voltage V_(DC) is higherthan the AC line voltage across an entire cycle of the AC line voltageas discussed in additional detail below.

First, the pump/flying capacitor (C_(P)) is preferably large enough tostore a maximum input charge from the AC mains, which input charge isfunction of the output power, the peak input voltage and the switchingfrequency of the power converter. Those constraints ensure the pumpcapacitor (C_(P)) can store the maximum charge needed. This maximumcharge occurs at the peak input current and voltage:

$\begin{matrix}{C_{P} \geq \frac{2P_{OUT}}{\eta \cdot f_{s} \cdot {V_{IN\_ pk}}^{2}}} & (12)\end{matrix}$

Second, the capacitance of the smoothing capacitor (C_(DC)) ispreferably dimensioned such that the voltage across it, which maycorrespond to the DC supply voltage (V_(DC)) of the resonant inverter ishigher than the AC line voltage, or mains input voltage, across theentire cycle of the AC line voltage at least during steady stateoperation of the power converter 100. That constraints ensures the ACrectification circuit (D_(B)) or diode bridge and the pump diode (D_(P))do not conduct at the same time. Hence, avoiding a direct current flowfrom the AC line voltage to the smoothing capacitor (C_(DC)) andtherefore providing control on the input current, which has to flowthrough the pump capacitor C_(p).

$\begin{matrix}{C_{DC} \geq \frac{P_{OUT}}{2{\omega_{l} \cdot V_{DC\_ ripple} \cdot V_{DC\_ avg}}}} & (13)\end{matrix}$

As mentioned before, the class-DE stage is preferably designed with ahigh voltage gain (close to unity) which markedly reduces the dependenceof the input current on the voltage across the smoothing capacitor(C_(DC)) and the output voltage and makes it a function of only the ACinput voltage under steady state operation of the power converter 100.

FIG. 5 shows low-frequency operation of the charge pump circuit 103,where the pump capacitor (C_(P)) gets charged and discharged within falland rise times of the rectifier input voltage V_(REC), respectively. Theelectrical charge ΔQ_(P) is largely proportional to a voltage differenceacross the pump capacitor C_(P) and this voltage difference variesbetween a low-frequency high-value V_(P_high) and a constant low-valueV_(P_low) as indicated on the waveform plot 505. The circuit ensuresthat the electrical charge variation of pump capacitor C_(P), which isproportional to the voltage variation across C_(P), i.e.V_(P_high)−V_(P_low), follows, or varies proportionally with, the ACline voltage V_(IN) across the line cycle 50/60 Hz. Accordingly, theaverage of the mains line current I_(IN) follows or tracks instantaneousamplitude of the AC mains line voltage with high accuracy andconsequently a power factor very close to unity can ideally be obtained.Experimental results described below illustrates that an impressivelyhigh power factor in the range 0.95-0.99 is readily obtainable.

The skilled person will appreciate that the rectifier input voltageV_(REC) can be any kind of waveform with a substantially constant ACamplitude and that any DC bias of the rectifier input voltage V_(REC)has no effect on the line input current shape. This feature providescompatibility with different arrangements of the resonant tank circuitincluding the parallel-resonant, LCC, and LLC tank topologies or circuitarrangements.

FIG. 6 illustrates high frequency operation across two switching cyclesof the switching frequency of the class DE resonant converter (105) andincludes inter alia waveforms of the charging current I_(P) and mainsline current I_(IN) showing the AC-DC converter operation at a maximumpower point (ω_(l)t=π/2) to illustrate the charge pump circuitoperation. The maximum power point corresponds to a resonant frequencyof the series resonant tank. Across every switching cycle, atsteady-state, the operation of the charge pump circuit spans over fourintervals 1, 2, 3 and 4 or states as indicated along the time axis ofthe waveform plots on FIG. 6, as follows:

1) In a first interval, the rectified line voltage V_(B) as shown onplot 621 is lower than the DC supply voltage V_(DC) and higher than theline input voltage V_(IN) where the V_(IN) waveform refers to thevoltage on the node interfacing the LC-mains filter and the diode bridgeD_(B) so both diodes are off and no current flows through the pumpcapacitor C_(P) and the voltage V_(P) as shown on plot 623 isessentially constant at V_(P_low).

2) A second interval or states takes place during the fall time of therectifier input voltage V_(REC) as shown on plot 622. Once V_(REC)starts to decrease, V_(B) has to decrease along until diode D_(B) getsforward biased and the rectified line voltage V_(B) gets pulled toV_(IN). While V_(REC) continues decreasing while V_(B) remainssubstantially constant, because the 50/60 Hz grid frequency varies muchslower, e.g. with a factor 1000 or more, than the switching frequency ofV_(REC), V_(P) increases and the pump capacitor C_(P) is charged by themains line current I_(IN) as shown on plot 627, until V_(REC) reachesits low-value and V_(P) reaches its high-value, where:

V _(P_high) =V _(IN) −V _(REC_low)  (20)

3) A third interval or state begins once V_(REC) settles at thelow-value, where C_(P) stops charging while diode D_(P) still remainsnon-conducting or blocking. Similar to the first interval, no currentflows through the pump capacitor and the voltage across the pumpcapacitor C_(P) remains substantially constant.

4) Eventually, a fourth interval or state takes place during a rise timeof V_(REC). Once V_(REC) starts to increase, V_(B) is forced increasealong until pump diode D_(P) gets forward biased or conducting and V_(B)is pulled to the DC supply voltage V_(DC). While V_(REC) (622) continuesincreasing, with DC supply voltage V_(DC) constant, V_(P) decreases andpump capacitor C_(P) deliver a discharge current into the smoothingcapacitor C_(DC) to increase the DC supply voltage V_(DC) and decreasethe voltage across pump capacitor C_(P) until V_(REC) reaches itshigh-value, V_(REC_high), and V_(P) reaches its low-value, where

V _(P_low) =V _(DC) −V _(REC_high)  (21)

By the end of the fourth interval, the operation of the charge pumpcircuit reverts to interval or state 1) again and the cycle repeats.

This analysis shows that the mains line current I_(IN) on plot 627 isdiscontinuous. The positive charging current I_(P) on current waveformplot 625 is likewise discontinuous and only flows into the charge pumpcircuit during the second interval or state. As illustrated by thecurrent waveform plot 627 of the mains line current I_(IN) on FIG. 6, acharging current pulse having the electrical charge ΔQ_(P) is drawn fromthe AC mains line voltage during every interval 2 period and theelectrical charge ΔQ_(P) varies substantially proportionally with theinstantaneous amplitude of the AC line voltage. As illustrated by thecurrent waveform plot 625 of the pump capacitor current I_(P) on FIG. 6,a corresponding charging current pulse, possessing an electrical chargeΔQ_(P), is drawn into the pump capacitor C_(P) during every interval 2)state. However, a corresponding charging current pulse is drawn out ofthe pump capacitor C_(P) during every interval 4) state and thatelectrical charge is supplied into the smoothing capacitor C_(DC) toincrease the DC supply voltage V_(DC).

FIG. 4 is a schematic electrical diagram of a first exemplary embodimentof the previously discussed AC-DC power converter 100 based on aclass-DE series resonant inverter as schematically illustrated on FIG.2. The AC-DC power converter 100 comprises an optional mains inputfilter including inductor L_(IN) and capacitor C_(IN) inserted betweenthe mains voltage and the AC rectification circuit 401 which maycomprise a diode bridge as illustrated. A class DE resonant inverter isconfigured to convert the DC supply voltage V_(DC) into the resonantinverter voltage V_(REC) at a fixed or adjustable switching frequency.The class DE resonant inverter 405 comprises high-side and low-sidesemiconductor switches Q_(HS) and Q_(LS) which are driven in anon-overlapping manner by gate driving circuit 413. The gate drivingcircuit 413 may comprise a digital isolator, such as Si8610BC by SiliconLabs, and an off-the-shelf type of gate driver such as UCC27611 by TexasInstruments, for each of the semiconductor switches Q_(HS) and Q_(LS).The output voltage V_(SW) of the half-bridge driver comprising high-sideand low-side semiconductor switches Q_(HS) and Q_(LS) is applied to aseries resonant tank 414 to produce a flow of resonant current I_(RES)through the tank and a resonant inverter voltage V_(REC) at the outputof the tank 414.

The switching frequency or duty cycle may be adjusted or controlled byan output current or output voltage regulation loop or mechanism. Theoutput current or output voltage regulation loop may comprise aswitching frequency controller 440 or alternatively a duty cyclecontroller, i.e. using on/off control of the converter. One input of thecontroller 440 may be coupled to the output voltage V_(OUT) whileanother input is coupled to a voltage or current reference generator(not shown) which sets a target output voltage V_(REF) or target outputcurrent of the AC-DC power converter 100. The frequency controller 440generates a switch control signal F_(SW) to an input of the gate driver413 to adjust the switching frequency of the class DE resonant inverter405 such that a target DC output voltage (V_(OUT)) or target DC outputcurrent is achieved as discussed in additional detail below inconnection with experimental results of a prototype AC-DC powerconverter. The AC-DC power converter 100 additionally comprises a highfrequency AC-DC rectification circuit 416 which is configured togenerate the DC output voltage (V_(OUT)) to the converter/output load(R_(L)) by tapping off energy/power supplied by the series resonant tank414 of the class DE inverter.

The AC-DC power converter 100 additionally comprises a charge pumpcircuit operating according to the principles discussed in connectionwith FIG. 3 above and which performs the previously discussedadvantageous power factor correction (PFC) of the AC-DC power converter100. The charge pump circuit comprises pump or flying capacitor C_(P), asmoothing capacitor C_(DC) and pump diode D_(P). The rectifier inputvoltage which is equal to V_(REC) is applied at the junction nodebetween output rectification diodes D_(R1) and D_(R2). The pump orflying capacitor C_(P) is connected between the rectified line voltageV_(B) and the rectifier input voltage V_(REC) which is equal to theresonant inverter voltage produced by the class DE resonant inverter.

Table I shows target specifications of the AC-DC power converter 100according to one embodiment:

TABLE I Parameter Specification Operational Voltage 230 V_(RMS) LineFrequency  50 Hz Output Power  50 W Power Factor >0.9

The charge pump circuit 103 (FIG. 3) is preferably designed such thatthe capacitance of the pump capacitor C_(P) is large enough to store amaximum input current coming from the offline mains voltage. From theanalysis above and considering that for a series-resonant converter, theresonant inverter voltage V_(REC) varies between V_(OUT) and 0 V and canbe evaluated as follows:

V _(P_high) =V _(IN) −V _(REC_low) =V _(IN)−0=V _(IN)  (11)

V _(P_low) =V _(DC) −V _(REC_high) =V _(DC) −V _(OUT)  (12)

The equations (11), (12) show that the high values for the voltageacross the pump capacitor C_(P) take the envelope of the input voltage,while the low values take the envelope of the difference between theresonant converter's input and output voltages, which can be consideredconstant in high frequency converters.

Across one switching cycle, the variation of charge in the pumpcapacitor is equal to:

ΔQ _(P) =C _(P) ·ΔV _(P) =C _(P)(V _(P) _(high) −V _(P) _(low) )=C_(P)(V _(IN) −V _(DC) +V _(OUT))  (13)

The pump capacitor current I_(P) when averaged across one switchingcycle is equal to:

$\begin{matrix}{l_{P} = {\frac{\Delta Q_{P}}{T_{s}} = {{{f_{s} \cdot \Delta}\; Q_{P}} = {f_{s} \cdot {C_{P}\left( {V_{IN} - V_{DC} + V_{OUT}} \right)}}}}} & (14)\end{matrix}$

Considering the class-DE inverter stage operates near resonance with ahigh gain close to 1, the difference between the DC supply voltageV_(DC) and V_(OUT) will be very small. Therefore, at steady state, for aconstant switching frequency, the pump capacitor current or chargingcurrent I_(P) and, accordingly mains line current I_(IN), becomeproportional to the input voltage, i.e. line voltage, of the AC-DCconverter, resulting in a high power factor and low THD. In that case,the maximum current through the pump capacitor (averaged over aswitching cycle) will be equal to the maximum input current, which canbe calculated as follows. Assuming 90% converter efficiency, the averageinput power is equal to:

$\begin{matrix}{P_{IN\_ avg} = {\frac{P_{OUT\_ avg}}{\eta} = {55.5\mspace{11mu} W}}} & (15)\end{matrix}$

Assuming a power factor of 1, the input power is the product of twosinusoids, resulting in a peak input power of:

P _(IN_max)=2P _(IN_avg)=111 W  (16)

Accordingly, the maximum mains line current I_(IN) or input current is:

$\begin{matrix}{I_{IN\_ max} = {\frac{P_{IN\_ max}}{V_{IN\_ max}} = {\frac{111\mspace{11mu} W}{325\mspace{11mu} V} = {{342{mA}} = I_{P\_ max}}}}} & (17)\end{matrix}$

By substituting those values in equation (14), the capacitance for thepump capacitor C_(P) can be calculated as follows:

$\begin{matrix}{C_{P} = {\frac{I_{P\_ max}}{f_{s} \cdot V_{IN\_ max}} = {\frac{342\mspace{11mu}{mA}}{1\mspace{11mu}{MHz}*325\mspace{11mu} V} = {{1.0}6\mspace{11mu}{nF}}}}} & (18)\end{matrix}$

The value for C_(P) can be adjusted to account for the DE inverter stagegain not being exactly 1 (V_(DC)−V_(OUT)≠0). In this design, a value of1.3 nF is chosen. Whereas the maximum voltage seen by the pump capacitorC_(P) is equal to:

ΔV _(P_max) =V _(IN_max)=325 V  (19)

Based on the analysis and calculations outlined above, the exemplaryembodiment of the present AC-DC power converter was simulated usingLTspice. The switching frequency is set to 1.04 MHz for the simulation.The output power delivered to the converter load is 51 W and an averageoutput voltage of 301V. The power factor is determined to about 0.99 andtotal harmonic distortion (THD) of the mains line current is 5.4%. FIG.7 shows a mains line voltage waveform 750 of 230 V_(RMS), 50 Hz inputtedto the AC-DC converter for the LTspice circuit simulation. A currentwaveform 755 of the corresponding the mains line current I_(IN) 7 drawnby the AC-DC converter is also shown. It is evident that that the mainsline current I_(IN) is largely in-phase with the mains line voltage anddistortion of the current waveform is relatively low.

Due to the charge pump circuit operation, the current in the resonanttank 414 peaks to the same value as pump capacitor current I_(P), whichtakes place at the peak of the input power (ω_(l)t=π/2, 3π/4). Thecurrent in the resonant tank 414 is further a function of the switchingfrequency, the output power, and a shape of the V_(REC) waveform. Thevalue is obtained from the simulation results, as shown in FIG. 8, whichindicates that the resonant inductor current peaks to about 1.7 A. Thisis also the maximum value of pump capacitor current I_(P) at the maximumoutput power of the AC-DC power converter.

Error! Reference source not found. below summarizes specifications forthe design of an exemplary inductor L_(RES) of the series resonant tank(214) which are obtained from the circuit analysis and simulationresults as described above.

TABLE II Parameter Specifications Inductance 156 μH Current Frequency  1MHz Sinusoid Current Amplitude 1.7 A

When handling high frequency AC currents, a key factor to the inductordesign is choosing the right core material. Several magnetic materials[6][7] are investigated and compared in terms of core losses at 1 MHz,as shown in Error! Reference source not found. FIG. 10, wherein the 3F46material (Ferroxcube) may be chosen as it shows the lowest core lossesat the design operating conditions.

The following equation was used to estimate the inductor core loss. Thepeak flux density in the core can be calculated from:

$\begin{matrix}{B_{pk} = {\frac{1}{N} \cdot \frac{I_{pk} \cdot L}{A_{e}}}} & (22)\end{matrix}$

where N is the number of turns, I_(pk) is the amplitude of the resonantcurrent, L is the inductance, and A_(e) is the effective core area.Considering that the core loss is a function of the peak flux densityfor a chosen material, the loss for a given number of turns and coresize can be estimated. The following calculation of the DC resistance ofthe windings gives an estimate of the winding losses.

The total cross-sectional area of the windings may be calculated from:

A _(c) =n _(wires) ·π·r _(wire) ²  (23)

where n_(wires) is the number of strands of Litz wire and r_(wire) isthe wire radius. The DC resistance may then calculated from:

$\begin{matrix}{R_{dc} = {\rho_{cu} \cdot \frac{M\; L\;{T \cdot N}}{A_{c}}}} & (24)\end{matrix}$

where ρ_(cu) is the copper resistivity and MLT is the mean length ofturn. For an EFD 25/13/9 core size, with two parallel layers of 20*0.05mm Litz wire, the DC resistance is 8.6 mΩ·N. Next, AC resistance of thewindings is calculated. The skin effect is negligible when using Litzwire at 1 MHz, but the proximity effect can have a significant influenceon the closely wound wires. Modelling the AC resistance to be threetimes larger than the DC resistance estimates the winding losses to

$\begin{matrix}{P_{cu} = {{R_{ac} \cdot I_{rms}^{2}} = {{3 \cdot R_{dc} \cdot \frac{I_{pk}^{2}}{2}} = {37\mspace{11mu}{{mW} \cdot N}}}}} & (25)\end{matrix}$

Based on these estimates, the inductor is designed with 52 turns, whichhelps distribute the losses evenly between the core and the windings andresults in acceptable total losses. An airgap of 1.2 mm, distributedacross the three legs of the core, adjusted the desired inductance.

FIG. 4A is a schematic electrical diagram of a second exemplaryembodiment of the previously discussed AC-DC power converter 100 a basedon class-DE resonant inverter. Compared to the previously discussedfirst exemplary embodiment of the AC-DC power converter on FIG. 4, thepresent AC-DC power converter 101 additionally comprises a transformerT1 (no denotation) with a certain voltage conversion ratio to step-up orstep-down the resonant inverter voltage V_(res). The transformer T1provides galvanic isolation between primary side circuitry and secondaryside circuitry of AC-DC power converter 100 a and is coupled between theresonant output voltage V_(res) and an input node 411 a of the outputrectification circuit 416 a connected to diodes D_(R1) and D_(R2). Asdiscussed above in connection with non-isolated AC-DC power converter100 a, the charge pump circuit comprises a pump capacitor C_(P), asmoothing capacitor C_(DC) and pump diode D_(P). The pump capacitorC_(P) is connected between the rectified line voltage V_(B) and theresonant inverter voltage V_(REC) produced by the class DE resonantinverter. Preferably, the electrical interconnection between therectified line voltage V_(B) and resonant inverter voltage V_(REC) isbased on capacitor only network, for example exclusively including thepump capacitor C_(P). This provides a compact and low-cost connectionthat may be based on standard, low cost and readily available capacitorsas discussed below. Hence, the electrical interconnection between therectified line voltage V_(B) and resonant inverter voltage V_(REC) istherefore preferably carried out without the use of any inductivecoupling or component from the isolation transformer, T, such as aseparate transformer winding, back to the rectified line voltage V_(B).

FIG. 9 shows inductor losses in the inductor L_(RES) of the seriesresonant tank (214) vs. number of turns. Based on these estimates,L_(RES) is preferably designed with about 52 turns, which helpsdistribute the losses evenly between the core and the windings andresults in acceptable total losses. An airgap of 1.2 mm, distributedacross the three legs of the core may be utilized to adjust the desiredinductance. An experimental prototype of the AC-DC converter wasimplemented and assembled on a printed circuit board. Because of thecharge pump circuit operation, high frequency AC current runs throughthe AC rectification circuit, or input bridge, The latter is thereforepreferably implemented using four fast recovery diodes as rectificationelements. With respect to selection of the high-side and low-sidesemiconductor switches Q_(HS) and Q_(LS) of the half-bridge 412 theinventors found gallium nitride FETs to show superior performancecompared to the silicon super-junction and silicon carbide counterparts.However, the skilled person will appreciate that semiconductor switchesmay be used for the purpose based on design constraints imposed on anyspecific embodiment of the present AC-DC power converter. For therectification circuit 416, including diodes D_(R1) and D_(R2), siliconcarbide Schottky diodes may be employed because they show higher energyefficiency and thermal stability over the silicon counterparts.

Error! Reference source not found. shows a breakdown of the incorporatedpower stage components of the experimental prototype AC-DC powerconverter.

TABLE III Component Simulated Prototype Type L_(IN) 100 μH 100 μHInductor SLF7045 C_(IN) 30 nF 2 * 15 nF Ceramic (C0G) Diode Bridge 4 *ESH1GMRSG Si Fast Recovery C_(DC) 10 μF 1 * 10 μF Electrolytic 3 * 0.1μF Ceramic (C0G) D_(P) RF201LAM4S Si Fast Recovery C_(P) 1.36 nF 2 * 680pF Ceramic (C0G) Q_(HS), Q_(LS) GS66502B GaN Switches L_(RES) 156 μH 152μH Custom design C_(RES) 188 pF 220 pF Ceramic (C0G) D_(R1), D_(R2)GB01SLT06214 SiC Schottky C_(OUT) 30 nF 2 * 15 nF Ceramic (C0G)

The experimental prototype AC-DC power converter was tested foroperation from a mains voltage of 230 V_(RMS) and running at converterswitching frequencies between 0.96 MHz and 1.04 MHz. The latterfrequency range lies within an inductive mode of operation for theresonant converter such that soft-switching operation, i.e. ZVS, of theclass DC inverter was achieved.

FIG. 11 (top plot) shows the measured output power (left scale) andconversion efficiency (right scale) across the operational frequencyrange from 960 kHz to 1040 kHz of the experimental prototype AC-DC powerconverter. The depicted measurement results illustrate how output powermodulation or control is achieved from 26 W to 50 W of output powerthrough control of the switching frequency. At the same time a peakefficiency of 87.9% at 1.04 MHz switching frequency is achieved.

FIG. 11 (bottom plot) shows measured PFC results of the experimentalprototype AC-DC power converter. The PFC results exhibit a peak powerfactor of 0.99 and minimum THD of 8.6% at an output power of 50 W andswitching frequency of 960 kHz. The measured data shows that operationat lower frequencies, close to resonance of the series resonant tankwith higher gain achieves higher power factor and lower THD. This isconsistent with the analysis given in the previous sections.

FIG. 12 shows measured harmonics distribution of the mains line currentat full-load operation and at half-load operation of the experimentalprototype AC-DC power converter where THD figures of 8.6% and 17.4% aremeasured, respectively. Since one of the potential applications for theproposed converter is the rectifier or rectification circuit or stage inLED drivers, the figure illustrates the harmonics magnitudes against theIEC 61000-3-2 standard class-C device limits [1][2], where the histogramshows that the measured harmonics magnitudes are well-within the limitsset by the IEC standard.

FIG. 13 shows line-frequency time domain waveforms of the experimentalprototype AC-DC power converter. The experimental prototype AC-DC powerconverter is delivering an output or load power of 12.8 W and has ameasured power factor of 0.99 and a THD of 9.1% for an average DC outputvoltage of 179 V with 20 V low-frequency ripple as illustrated by theV_(OUT) waveform.

FIG. 14 shows measured harmonics distribution of the mains line currentfor the experimental prototype AC-DC power converter at 120 V_RMS and230 V_RMS line voltage inputs against the IEC 61000-3-2 standard class-Cdevice limits [1][2]. FIGS. 13 and 14 illustrate that a high powerfactor and low THD of the prototype AC-DC power converter are achievedwith different line input voltages as the charge pump circuit works inthe same manner.

REFERENCES

-   [1] IEC 61000-3-2, Fifth Edition, International Electrotechnical    Commission, 2018.-   [2] EN 61000-3-2, European Committee for Electrotechnical    Standardization, 2014.-   [3] X. Xie, C. Zhao, L. Zheng and S. Liu, “An Improved Buck PFC    Converter With High Power Factor,” in IEEE Transactions on Power    Electronics, vol. 28, no. 5, pp. 2277-2284, May 2013.-   [4] X. Wu, J. Yang, J. Zhang and M. Xu, “Design Considerations of    Soft-Switched Buck PFC Converter With Constant On-Time (COT)    Control,” in IEEE Transactions on Power Electronics, vol. 26, no.    11, pp. 3144-3152, November 2011.-   [5] D. C. Marian, K. Kazimierczuk, Resonant Power Converters, 2nd    Edition ed., Wiley-IEEE Press, 2011.-   [6] Ferroxcube material    datasheet—https://www.ferroxcube.com/upload/media/design/FXCStainmetzCoefficients.xls.-   [7] Micrometals material    datasheet—https://micrometalsarnoldpowdercores.com/pdf/mix/Mix-6-DataSheet.pdf.

1. An AC-DC power converter comprising: an AC rectification circuitconfigured to convert an AC line voltage into a rectified line voltage,a rectifying element connected between the rectified line voltage and aDC supply voltage of a resonant DC converter; said resonant DC convertercomprising: a resonant inverter configured to convert the DC supplyvoltage into a resonant inverter voltage at a fixed or controllableswitching frequency, and an output rectification circuit configured togenerate a DC output voltage or DC output current from the resonantvoltage for supply to a converter load, a charge pump circuit connectedto the rectified line voltage and the resonant inverter voltage, wheresaid charge pump circuit is configured to draw current pulses at theswitching frequency from the AC line voltage, wherein electrical chargesof the current pulses vary proportionally to instantaneous amplitude ofthe AC line voltage.
 2. An AC-DC power converter according to claim 1,wherein the charge pump circuit comprises: a smoothing capacitorconnected to the DC supply voltage of the resonant DC-DC converter; apump or flying capacitor connected from the resonant inverter voltage tothe rectified line voltage.
 3. An AC-DC power converter according toclaim 2, wherein the charge pump circuit is configured to, during acycle of the switching frequency, sequentially cycle through states of:a first state where each of the rectifying element and AC rectificationcircuit is non-conducting/off and a voltage across the pump/flyingcapacitor remains substantially constant; a second state where ACrectification circuit is conducting/on and the rectifying element isnon-conducting/off to charge the pump/flying capacitor by line currentdrawn from the AC line voltage; a third state where each of therectifying element and AC rectification circuit is non-conducting/offand the voltage across the pump/flying capacitor remains substantiallyconstant; a fourth state where the AC rectification circuit is in anon-conducting/off state and the rectifying element is in aconducting/on state such that discharge current flows from thepump/flying capacitor into the smoothing capacitor to increase the DCsupply voltage of the resonant inverter and decrease the voltage acrossthe pump/flying capacitor.
 4. An AC-DC power converter according toclaim 3, wherein the charge pump circuit is configured to: cycle throughits second state during a rising edge of a waveform of the resonantinverter voltage; and cycle through its fourth state during a fallingedge of the waveform of the resonant inverter voltage.
 5. An AC-DC powerconverter according to claim 2, wherein a capacitance of the smoothingcapacitor and a capacitance of the pump/flying capacitor are selectedsuch that the DC supply voltage of the resonant inverter is higher thanthe AC line voltage across an entire cycle of the AC line voltage.
 6. AnAC-DC power converter according to claim 1, further comprising: avoltage or current regulation loop configured to adjust the DC outputvoltage or DC output current in accordance with a DC reference voltageor a DC reference current.
 7. An AC-DC power converter according toclaim 6, wherein the voltage or current regulation loop is configuredto: adjust the DC output voltage or DC output current by adjusting thecontrollable switching frequency, and/or adjust the DC output voltage orDC output current by off/on duty cycle modulation of the fixed orcontrollable switching frequency.
 8. An AC-DC power converter accordingto claim 1, where in the resonant inverter comprises at least onesemiconductor switch connected between the DC supply voltage and anegative supply rail; said at least one semiconductor switch comprisingone or more wide bandgap transistors such as one or more gallium nitrideFET(s).
 9. An AC-DC power converter according to claim 8, wherein aswitch signal, at the fixed or controllable switching frequency, isapplied to a control terminal of the least one semiconductor switch ofthe resonant inverter.
 10. An AC-DC power converter according to claim1, where in the resonant inverter and/or the output rectificationcircuit of DC-DC converter is configured for zero voltage switching(ZVS) and/or zero current switching (ZVS).
 11. An AC-DC power converteraccording to claim 1, wherein the output rectification circuit comprisesone or more passive diodes, or one more active/controllable diodes. 12.An AC-DC power converter according to claim 1, where in the resonantinverter comprises a series resonant network and/or a parallel resonantnetwork having a predetermined resonance frequency.
 13. An AC-DC powerconverter according to claim 1, where in the switching frequency isbetween 100 kHz and 300 MHz.
 14. An AC-DC power converter according toclaim 1, further comprising galvanic isolation barrier coupled betweenthe resonant output voltage and an input of the output rectificationcircuit.
 15. A method of applying power factor correction to an AC-DCpower converter using a charge pump circuit, said method comprisingsteps of: converting an AC line voltage into a rectified line voltage,applying the rectified line voltage to a DC supply voltage of a resonantDC-DC converter through a rectifying element, generating a resonantinverter voltage by switching a resonant inverter at a fixed orcontrollable switching frequency, rectifying the resonant invertervoltage to generate a DC output voltage or DC output current, drawingcharging pulses, at the switching frequency, from the AC line voltageinto a pump or flying capacitor connected between the rectified linevoltage and the resonant inverter voltage, wherein electrical charges ofthe charging pulses vary proportionally to an instantaneous amplitude ofthe AC line voltage, discharging the pump or flying capacitor into asmoothing capacitor, connected to the DC supply voltage, by supplyingcurrent pulses, at the switching frequency, into the smoothingcapacitor.